1. Field of the Invention
The present invention generally relates to controlled threshold type electric devices and comparators employing the same. More particularly, the present invention relates to a controlled threshold type electric device which controls the threshold value of a transistor or the input threshold value of an amplifier through feedback control such that they converge to their desired values The present invention further relates to a comparator which compares, using such a controlled threshold type electric device, a reference input voltage and an external input voltage and outputs the result of comparison.
2. Description of the Background Art
A threshold voltage V.sub.T of an MOS (Metal Oxide Silicon) transistor or the like generally fluctuates depending on various parameters such as an initial threshold voltage V.sub.TO or substrate effect constant .gamma. which vary between different substrates. For example, the initial threshold voltage V.sub.TO may deviate from a desired value generally by about 100 mV to several 100 mV due to a slight difference of manufacturing conditions. Since accurate control of such parameters is difficult in manufacturing transistors, the threshold voltage V.sub.T requires some control based on measurements after the manufacturing to attain a desired value. To meet such requirement, for example, the technique described in U.S. Pat. No. 3,657,575 is generally known. FIG. 20 is a circuit diagram showing the controlled threshold type electric device described in this U.S. Pat. No. 3,657,575. Referring to the diagram, the conventional threshold controlled type electric device comprises a plurality of MOS transistors T.sub.1 to T.sub.4 and 1 and a control voltage generating means 40. The control voltage generating means 40 comprises a bipolar transistor 2, a diode 3 and resistors 5 and 4. The transistor 1 receives a reference voltage E.sub.R from a DC source 6 at its frontgate G. At its substrate as an example of the backgate, the transistor 1 receives a control voltage V.sub.2 from the control voltage generating means 40. As a result, a backgate bias voltage V.sub.B is generated between the frontgate G and the backgate of the transistor 1. This backgate bias voltage V.sub.B can control the threshold of the transistor 1. The control voltage generating means 40 receives an operating voltage Vd which causes generation of a feedback input voltage V.sub.1 at a terminal 48 and also generation of a drain current I.sub.D and a base current I.sub.B. Assuming that the base current I.sub.B is substantially smaller than the drain current I.sub.D, the feedback input voltage V.sub.1 is represented as V.sub.1 =V.sub.d -I.sub.D .times.R.sub.L, where R.sub.L is resistance value of the resistor 4. The drain current I.sub.D varies depending on magnitude of the threshold value of the transistor 1, which varies in turn depending on magnitude of the backgate bias voltage V.sub.B. Further, the backgate bias voltage V.sub.B varies depending on magnitude of the control voltage V.sub.2 generated by the control voltage generating means 40. A change in the drain current I.sub.D leads to a change in the feedback input voltage V.sub.1. Since the feedback input voltage V.sub.1 is applied to the base of the bipolar transistor 2 through the diode 3, the change in the feedback input voltage V.sub.1 causes the control voltage V.sub.2 to change correspondingly. Thus, a negative feedback loop is configured of the transistor 1 and the control voltage generating means 40 in which the threshold value of the transistor 1 is used as a controlled variable, the voltage V.sub.1 generated at the terminal 48 as a feedback signal, and the control voltage V.sub.2 as a control action signal. As a function of the negative feedback loop, the control voltage V.sub.2 is applied to the backgate of the transistor 1 so that the threshold value of the transistor 1 may converge to its desired value, and the same control voltage V.sub.2 is applied to the backgates of the other transistors T.sub.1 to T.sub.4. As a result, also those other transistors T.sub.1 to T.sub.4 are controlled such that their threshold values converge to the respective desired values in the same manner as the transistor 1. Consequently, deviation of the threshold values in manufacturing or that caused by a temperature change and the like in use can be prevented, keeping the threshold values of transistors at their desired values.
In the conventional threshold controlled type electric device shown in FIG. 20, the control voltage generating means 40 comprises the bipolar transistor 2. When a Bi. MOS process (bipolar element-MOS mixed process) is employed to obtain the bipolar element in manufacturing the controlled threshold type electric device, it will result in an increased cost. Meanwhile, though MOS standard process can not basically provide bipolar transistors, CMOS process can exceptionally provide them as parasitic elements. Since this CMOS process has not been optimized in itself for the production of bipolar transistors, the resulting bipolar transistors have low performance and require a considerable base current for operation. If the controlled threshold type electric device shown in FIG. 20 is realized using such an element, the base current I.sub.B take a relatively large value with respect to the drain current I.sub.D, causing the following problems.
When a voltage drop by the diode 3 is shown as V.sub.D, that by the bipolar transistor 2 as V.sub.BE, and a threshold value corresponding to the backgate voltage V.sub.2 (a function of V.sub.2) as V.sub.TH (V.sub.2), the expression of voltage drop along the resistor 4 having a resistance value R.sub.L will be given as follows. EQU R.sub.L (I.sub.B +I.sub.D)=V.sub.d -(V.sub.2 +V.sub.BE +V.sub.D) (1)
The expression showing the drain current of the transistor 1 is as follows: ##EQU1##
where .beta. is a proportional constant. Therefore, when the drain current obtained from the expression (1) and that given by the expression (2) coincide with each other, the converged value of the threshold voltage V.sub.TH will satisfy the following expression. ##EQU2## As described in the U.S. Pat. No. 3,657,575, only when the base current I.sub.V is small enough and the circuit has an amplifying function, that is, when R.sub.L is large enough, the left-side of the expression (3) becomes 0, and the threshold value V.sub.TH (V.sub.2) meeting this relationship coincides with E.sub.R. In other cases, however, V.sub.TH (V.sub.2) does not coincide with E.sub.R. Particularly when the base current I.sub.B is large, V.sub.2 converges to a certain value irrespective of the transistor 1 being in the off-state. That is, V.sub.2 converges to a value meeting the expression below irrespectively of E.sub.R. ##EQU3## As a result, there arises the problem that the threshold value V.sub.TH can not be controlled by the reference voltage E.sub.R. The foregoing will be explained with reference to FIG. 21. FIG. 21 is a diagram for explaining occurrence of such a state where the reference voltage E.sub.R can not control the threshold value V.sub.TH. In FIG. 21, the ordinate represents the drain current I.sub.d and the abscissa represents the control voltage V.sub.2. A curve of solid line represents the drain current I.sub.d shown by the expression (2), a dashed line represents the drain current I.sub.d given by the expression (1) in the case of a large R.sub.L and a small I.sub.B, and a two-dotted chain line represents the drain current I.sub.d given by the expression (1) in the case of a large I.sub.B. When R.sub.L is large and I.sub.B is small as shown by the dashed line in FIG. 21, a desired value DV1 of the threshold value V.sub.TH coincides with the reference voltage E.sub.R. When I.sub.B is large as shown by the two dotted chain line, however, a desired value DV.sub.2 of the threshold value V.sub.TH does not coincide with the reference voltage E.sub.R. Consequently, the conventional controlled threshold type electric device requires a high-performance bipolar transistor which can not be manufactured through the conventional MOS standard process.
As another technique for making the input threshold voltage of an inverting amplifier or the like converge to a desired value, using feedback control, to prevent deviation of the input threshold value in manufacturing process or that caused by temperature and the like in use, and thus keeping the input threshold value always at the desired value, a controlled threshold type electric devices such as described in U.S. Pat. No. 4,791,318 and International Laying-Open Gazette of PCT application (International Laying-Open No. WO83/00785) is generally known. Such a conventional controlled threshold type electric device comprises two inverting amplifiers and one differential amplifier, in which the differential amplifier receives a voltage corresponding to the input threshold value of one inverting amplifier as a feedback input voltage and receives also a reference input voltage to compare the feedback input voltage and the reference input voltage, and then outputs such a control voltage as making the difference 0 to the inverting amplifier. That is, the one inverting amplifier and the differential amplifier constitute a negative feedback loop which exercises feedback control such that the input threshold value of the one inverting amplifier converges to its desired value, using the input threshold value of the one inverting amplifier as a controlled variable, the control voltage outputted from the differential amplifier as a control action signal, and a voltage corresponding to the input threshold value of the one inverting amplifier as a feedback signal. Through this feedback control, the control voltage is outputted from the differential amplifier which makes the input threshold value of the one inverting amplifier converge to its desired value (corresponding to the reference input voltage V.sub.1). The same control voltage is applied also to the other inverting amplifier so that the input threshold value of the inverting amplifier is controlled likewise to attain its desired value.
In these conventional electric devices for controlling input threshold values, each inverting amplifier is formed of a series connection of voltage controlled current sources each comprising a single transistor or parallel-connected transistors, so that the amplification factor of the inverting amplifier does not amount to a large value. In the following, description will be made on the amplification factor of a general inverting amplifier.
Generally, a transistor itself functions as a voltage controlled current source. Thus, a change in the gate-source voltage causes a change in the drain current. If .sub..DELTA. V.sub.G, .sub..DELTA. V.sub.S and .sub..DELTA. I.sub.D1 are change amounts in the gate potential, source potential and drain current, respectively, as shown in FIG. 15, the following relation is obtained: EQU .sub..DELTA. I.sub.D1 =gm (.sub..DELTA. V.sub.G -.sub..DELTA. V.sub.S)
where gm is a proportional constant. If the drain-source voltage is changed, the drain current is also changed although slightly, due to a channel length modulation effect or the like. If the change amount in the drain current due to such effect is represented as .sub..DELTA. I.sub.D2 the following relation is obtained: EQU .sub..DELTA. I.sub.D2 =go (.sub..DELTA. V.sub.D -.sub..DELTA. V.sub.S)
where .sub..DELTA. V.sub.D is the change amount in the drain voltage and go is the proportionality constant. Accordingly, a total change amount .sub..DELTA. I.sub.D in the drain current is expressed as follows: EQU .sub..DELTA. I.sub.D =.sub..DELTA. I.sub.D1 +.sub..DELTA. I.sub.D2 =gm (.sub..DELTA. V.sub.G -.sub..DELTA. V.sub.S) +go (.sub..DELTA. V.sub.D -.sub..DELTA. V.sub.S) (4)
However, since the change amount of V.sub.GS contributes more to the change amount of I.sub.D than that of V.sub.DS as described above, the following relation is established: EQU gm&gt;&gt;go (5)
The above description is related with respect to N-channel MOS transistors. The same phenomenon occurs also in P-channel MOS transistors. In the latter case, if the direction of the drain current is defined as shown in FIG. 16, the following expression is obtained: EQU .sub..DELTA. I.sub.D =-gm (.sub..DELTA. V.sub.G -.sub..DELTA. V.sub.S)-go (.sub..DELTA. V.sub.D -.sub..DELTA. V.sub.S) (6)
Next, let us assume a case in which transistors are connected in series as shown in FIG. 17. In this case, certain voltages (V.sub.B and V.sub.R) are applied to the gate of the transistor Q.sub.2 and the source of the transistor Q.sub.1. The change amount .sub..DELTA. I.sub.D1 of the drain current of the transistor Q.sub.1 is represented as follows: ##EQU4##
Therefore, the following equation is obtained: EQU .sub..DELTA. I.sub.D1 =gm1 .sub..DELTA. V.sub.G +go1 .sub..DELTA. V.sub.X ( 7)
The change amount .sub..DELTA. I.sub.D2 of the drain current of the transistor Q.sub.2 is represented as follows: ##EQU5##
Therefore, the following equation is obtained: EQU .sub..DELTA. I.sub.D2 =-gm2.sub.66 V.sub.X +go2(.sub..DELTA. V.sub.out -.sub..DELTA. V.sub.X) (8)
Since the change amounts of the drain currents of both transistors Q.sub.1 and Q.sub.2 are coincident with .sub..DELTA. Iout, the following equations are obtained from those equations (7) and (8). ##EQU6##
If .sub..DELTA. V.sub.X is eliminated from the equations (9) and (10), the following equation is obtained. ##EQU7##
If go1 and go2 are disregarded with respect to gm1 and gm2 from the equation (5), the following equation is obtained: ##EQU8## Comparing the equation (7) in the case of the transistor Q1 as a single device and the equation (11) in the case of using the transistors Q1 and Q2, the change amount of Iout with respect to the change amount of V.sub.G is the same and the change amount of Iout with respect to the change amount of Vout (or V.sub.X) is suppressed.
Thus, from the expression (5), the following expression is obtained. ##EQU9## Similarly, in the case of using P-channel MOS transistors, the following relation is obtained as shown in FIG. 18.
The inverting amplifiers in the conventional controlled-threshold type electric device as described above are formed by series connection of voltage controlled current sources 40 and 42 and in view of the conductivity of each transistor, they are shown as in FIG. 19.
However, the change amounts .sub..DELTA. I.sub.1, .sub..DELTA. I.sub.2 of I.sub.1, I.sub.2 are expressed by the following equations. EQU .sub..DELTA. I.sub.1 =Gm.sub.1.DELTA. V.sub.2 +Go.sub.1.DELTA. Vout EQU .sub..DELTA. I.sub.2 =Gm.sub.2.DELTA. Vo-Go.sub.2.DELTA. Vout
where Gm.sub.1, Gm.sub.2, Go.sub.1 and Go.sub.2 are constants having the same characteristics as gm.sub.1, gm.sub.2, go.sub.1 and go.sub.2, respectively.
Meanwhile, since Vo becomes constant after setting of the circuit, .sub..DELTA. Vo=0.
Since I.sub.1 becomes equal to .sub..DELTA. I.sub.2, the following equation is obtained. EQU GM.sub.1.DELTA. V.sub.2 +Go.sub.1.DELTA. Vout=-Go.sub.2.DELTA. Vout
Accordingly, the amplification factor AN is expressed as follows. ##EQU10##
Now, since the change amount of Iout (Go1 or Go2) with respect to that of Vout of a single transistor is relatively large, the amplification factor AN of an inverting amplifier in the above-described conventional electric device for controlling input threshold value does not become so large.
FIG. 7 is a circuit diagram showing a general A/D converter. The A/D converter comprises a plurality of comparators 500 which compare a reference input voltage with an external input voltage to output the comparison result. The controlled threshold type electric device as described in the above-mentioned International Laying-Open Gazette of PCT Application (International Laying-Open No. WO83/00785) is applied to each of the comparators 500. As previously described, however, since this conventional controlled threshold type electric device does not have a sufficient amplification factor, that is, amplitude of the output voltage Vout, when applied to a comparator of the A/D converter, the device is subject to malfunctions due to noise and the like, and may prevent precise conversion from analog to digital.